1. Technical Field
The present disclosure relates to a control device for a power factor correction device in forced switching power supplies.
2. Description of the Related Art
The use of devices is generally known for active correction of the power factor (PFC) for forced-switching power supplies employed in commonly used electronic apparatus such as computers, television sets, monitors, etc., and for supplying fluorescent lamps, i.e., stages of forced-switching pre-regulators that have the task of absorbing from the line a current that is almost sinusoidal and in phase with line voltage. Thus a forced-switching power supply unit of the current type comprises a PFC and a converter of continuous current into continuous current or DC-DC converter connected to the output of the PFC.
A forced-switching power supply unit of traditional type comprises a DC-DC converter and an input stage connected to the electric energy distribution line constituted by a full-wave diode rectifier bridge and by a capacitor connected immediately downstream so as to produce non-regulated continuous voltage from the alternating sinusoidal line voltage. The capacitor has sufficient capacity for the terminals thereof to have relatively small ripple with respect to a direct level. The rectifying diodes of the bridge will therefore conduct only for a small portion of each half cycle of the line voltage, as the instantaneous value of the latter is less than the voltage on the capacitor for the greater part of the cycle. As a result, the current absorbed by the line will consist of a series of narrow pulses the width of which is 5-10 times the resulting average value.
This has significant consequences: the current absorbed by the line has peak and root-mean-square (RMS) values that are much greater than in the case of absorption of sinusoidal current, line voltage is distorted through the effect of the pulsed absorption that is almost simultaneous with all the installations connected to the line, in the case of three-phase systems the current in the neutral conductor is greatly increased and there is little use of the energy potential of the electric-energy production system. In fact, the waveform of impulsive current is very rich in uneven harmonics that, although they do not contribute to the power delivered to the load, contribute to increasing the effective current absorbed from the line and therefore to increasing the dissipation of energy.
In quantitative terms, all this can be expressed both in terms of Power Factor (PF), defined as the ratio between the real power (the power that the power supply unit delivers to the load plus the power dissipated therein in the form of heat) and the apparent power (the product of the effective line voltage by the effective current absorbed) both in terms of Total Harmonic Distortion (THD), generally understood to be a percentage ratio between the energy associated with all the higher order harmonics and that associated to the fundamental harmonic. Typically, a power supply unit with a capacitive filter has a PF comprised between 0.4-0.6 and a THD greater than 100%.
A PFC arranged between the rectifier bridge and the input to the DC-DC converter enables a current to be absorbed from the line that is almost sinusoidal and in phase with the voltage, making the PF near 1 and reducing the THD. In order for the boost converter to operate correctly, the output voltage generated must always be greater than the input voltage. In the most typical embodiment thereof, in a PFC pre-regulator the output voltage is fixed around 400V in such a way as to be greater than the line peak voltage along the entire variation interval thereof (from 124.5 to 373.4 V in the case of a universal supply). In another embodiment, that of the so-called “boost follower” or “tracking boost”, the output voltage is set at a value that depends on the effective input voltage, but which is nevertheless greater than peak voltage.
Alongside the two traditional control methods of a PFC pre-regulator, i.e., pulse width modulation (PWM) at fixed frequency (FF) of “average current-mode” type with continuous conduction of current into the inductor (CCM) suitable for high power, and variable frequency PWM control of “peak current-mode” type, is a “Transition Mode” (TM) in which the system works at the border between continuous current mode (CCM) and discontinuous current mode (DCM) for conducting current into the inductor, suitable for lower power levels. Recently, the so-called “constant Toff control” or “Fixed-Off-Time” (FOT) control has had growing success, where Toff is the switch-off time of the power transistor. The reason for the interest in this method, especially during the critical power band (from 150 to 350 W), where the selection between TM and FF-CCM control types is often complex, is due to the fact that it combines the simplicity and low cost of the TM approach with the capacity to carry power (or the best current form factor) and the low content of radio frequency injected into the energy distribution line of the CCM/FF approach.
The FOT methodology consists of using the “peak current-mode” type control, like that of the TM systems, and of controlling the power switch of the converter so that in each switching cycle it remains switched off for a fixed time and the feedback used to regulate the output voltage of the PFC operates only on the duration of the switch-on of the switch.
In FIG. 1 there is shown schematically a constant PFC to Toff pre-regulating phase comprising a boost converter 20 and a control device 1. The boost converter 20 comprises a full wave rectifier bridge 2 having in input an alternating line voltage Vin, a capacitor C1 (that acts as a filter for the high frequency) having the terminals connected to the terminals of the diode bridge 2, an inductance L connected to a terminal of the capacitor C1, a power MOS transistor M having the drain terminal connected to a terminal of the inductance L downstream of the latter and having the source terminal coupled with ground by means of a resistance Rs suitable for enabling the current to be read that flows in the transistor M, a diode D having the anode connected to the common terminal of the inductance L and of the transistor M and the cathode connected to a capacitor Co having the other terminal connected to ground. The boost converter 20 generates a direct output voltage Vout on the capacitor Co that is the input voltage of a user stage that is cascade-connected, e.g., a DC-DC converter.
The control device 1 has to maintain the output voltage Vout at a constant value by means of a feedback control action. The control device 1 comprises an operational error amplifier 3 that is suitable for comparing part of the output voltage Vout, i.e., the voltage Vr given by Vr=R2*Vout/(R2+R1) (where the resistances R1 and R2 are serially connected to one another and are connected in parallel to the capacitor Co) with a reference voltage Vref, for example of the value 2.5V, and suitable for generating an error signal Se that is proportional to the difference between them. The output voltage Vout has a ripple and a frequency that is twice that of the line and is imposed on the continuous value. If, nevertheless, the bandwidth of the error amplifier is reduced significantly, (typically below 20 Hz) by means of the use of a suitable compensating network comprising at least a capacitor and having almost stationary operation, i.e., with effective input voltage and output load that are constant, this ripple will be greatly attenuated and the error signal will become constant.
The error signal Se is sent to a multiplier 4 where it is multiplied by a signal Vi given by a part of the line voltage rectified by the diode bridge 2. At the output of the multiplier 4 there is present a signal Imolt given by a rectified sinusoid, the width of which depends on the effective line voltage and on the error signal Se. Said signal Imolt represents the sinusoidal reference for the modulation PWM. Said signal is an input signal into the non-inverting terminal of a comparator 6 at the inverting input of which there is the voltage on the resistance Rs that is proportional to the current IL.
If the input signals entering the comparator 6 are equal the comparator 6 sends a signal to a control block 10 that is suitable for driving the transistor M and which in this case switches it off; so the output of the multiplier produces the peak current of the MOS transistor M that is enveloped by a rectified sinusoid. The block 10 comprises a set-reset flip-flop 11 having the reset input R that is the output signal from the comparator 6, the input set S, that is an output signal from a timer 13 and having an output signal Q. The signal Q is sent as an input to a driver 12 that commands switching on or off of the transistor M. The signal Q activates the timer 13, which after a preset period of time Toff has elapsed, sends a pulse to the input set S of the flip-flop 11 causing the transistor M to switch on. The period of time Toff can be modified from the exterior using a controller 14.
During the period of time Toff in which the transistor M is switched off the inductor L discharges the energy stored therein onto the load. If the time Toff is sufficient to discharge completely the inductor L in that switching cycle, operation will be of DCM type, otherwise operation will be of the CCM type.
The current absorbed from the line will be the low-frequency component of the current of the inductor L, i.e., the average current per switching cycle (the switching frequency component is almost totally eliminated by the line filter located at the input of the boost converter stage, which is always present in compliance with electromagnetic compatibility regulations). As the inductor current is enveloped by a sinusoid, low-frequency currency will have a sinusoidal trend. The control acts by modulating the duration of the switched-on period Ton but maintaining the switch-off period Toff constant, so that the operating frequency of the pre-regulator will vary from cycle to cycle according to the variation of the alternating line voltage, in particular, it varies in function of senθ with θ being the phase angle of the alternating line voltage.